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AND8031/D Isolated Precision Regulation of a Single 1.8 Volt Output from a Universal Line Input
Prepared by: Jason Hansen ON Semiconductor
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APPLICATION NOTE
Generic Off-Line Conversion Circuit
INTRODUCTION The following Application Note describes an off-line switching power supply utilizing a precision programmable reference to regulate a 1.8 volt output. The center of the app note is the MC33363B, a monolithic SMPS controller with a 700 volt power switch, and the NCP100, a sub-one volt precision programmable reference. The system design and analysis will be described in detail. The design requirements are for a universal off-line converter with a 1.8 volt, 1.0 ampere single output with less than 50 millivolts ripple and operates at 100 kHz. Most of the components selected are surface mount. This design is separated into the generic circuit of the off-line converter and the feedback network.
D1 R1 4.7 Universal Input C1 1n R2 4.7 R4 3.9 k 4.7 400 V C6 10 C5 47 p
To reduce the total number of components and thus minimize circuit board area, an SMPS controller with an integrated power switch is selected. With the low output power requirement, the MC33363B is selected as the control and power switch IC. The MC33363B contains an externally programmable frequency and current limit, internal startup circuit and can handle up to 8 watts of output power. The basic off-line conversion circuit is illustrated in Figure 1. The line input is filtered through the EMC circuit then rectified and filtered. The rectified voltage is converted to a lower voltage via the transformer and the MC33363B. The secondary of the transformer is rectified to a DC voltage and filtered. The output voltage controls the duty cycle of the switcher via the isolated feedback network. The calculated values are a starting point and do not replace bench testing.
U2 CTX22-14966 D5 MURS160 D7 MBRD835L + C7 820 D6 MURS120 R12 120 R9 12 k C8 820 C12 1 1.8 V -
D2
D3
D4 1N4005 1 3 4 5 R3 30 k 6 C3 390 p 7 C4 1 8
U1 16
Q1 BC858ALT1 C10 1 R11 10
C9 0.1
13 12 11 10 9 R6 2.7 k
U4 NCP100
R10 7.5 k
U3 SFH615-4
MC33363B
Figure 1. MC33363B Basic Flyback Circuit Schematic
(c) Semiconductor Components Industries, LLC, 2000
1
October, 2000 - Rev. 1
Publication Order Number: AND8031/D
AND8031/D
Since the power level is low, an RC EMC filter is utilized. The RC filter is not as efficient as an LC, but it uses less board area and will cost less. 1N4005 600 volt fast diodes provide the full wave rectification of the AC input. The bulk capacitor is the filter. The selection of the capacitor is determined from three factors: input voltage, ripple current and maximum output ripple. The equations for each are as follows.
Vdc + Vac * 2 Irms + Cin + Ipk2 * D 3
(Discontinuous mode only)
Psec + (Vout ) Vdiode) * Iout
1
(eq. 5) (eq. 6)
Pin + 2
1
(Vdc * Ton)2 , or Lpri * T (Vdc * Ton)2 Pin * T 2*P* T L
Lpri + 2
(eq. 7) (eq. 8)
Ipeak +
(eq. 1) (eq. 2)
Lpri Np 2 + Lsec Ns
(eq. 9)
k * Pin (AV) f * (Vripple (p-p))2 Pin + Pout h
(eq. 3) (eq. 4)
Where k is 1 for AC inputs the peak to peak ripple is 6.0 volts and is estimated to be 70% with the low power level of the board and the RC input filter. Solving for the equations above and using 50% for the maximum duty cycle, Vdc maximum is 375 volts, Vdc minimum is 120 volts, Ipeak and Irms are solved for in the transformer calculations, Pin is 3.2 watts, and Cbulk is 0.9 uF.(1) The transformer converts the rectified line voltage to the output voltage and is controlled by the MC33363B. To design the transformer, the following data is necessary. Vdc minimum is 100 volts (allowing for bulk voltage ripple and power switch voltage drop), the frequency is 100 kHz, the maximum duty cycle of the power switch and the reset time for the secondary are both 45% to maintain discontinuous conduction, and the secondary diode forward voltage drop is 0.45 volts (estimated).
For the primary side of the transformer, Lpri is 3.16 mH, Ipeak is 142 milliamperes, ton is 4.5 usec and Irms is 55 milliamperes-rms. For the secondary, Psec is 2.25 watts, L is 2.28 uH, Ipeak is 4.44 amperes and Irms is 1.72 amperes-rms. The turns ratio is 37.2. The auxiliary winding is set to 12 volts and 10 milliamperes. Using the secondary turns ratio, the auxiliary turns ratio is calculated as 6.6 in reference to the primary. With the peak primary current and the frequency determined, the components surrounding the MC33363B can be selected. Referring to Figures 2 and 3, RT will be set to 30 k Ohms and CT will be set to 390 pF. The Over Voltage Protection, pin 11, will not be used. Its main function is for loss of optocoupler protection. Pin 8, the reference voltage, requires a 1.0 uF ceramic capacitor for stability. The voltage compensation and voltage feedback, pins 9 and 10, will be covered in the feedback section of this document. The Vcc pin requires a 10 uF capacitor for stability.
fOSC, OSCILLATOR FREQUENCY (Hz)
CT = 100 pF
500 k C = 200 pF T 200 k CT = 500 pF 100 k 50 k 20 k
CT = 1.0 nF CT = 2.0 nF CT = 5.0 nF C = 10 nF
VCC = 20 V TA = 25C
IPK, POWER SWITCH PEAK DRAIN CURRENT (A)
1.0 M
1.0 0.8 0.6 0.4 0.3 0.2 Inductor supply voltage and inductance value are adjusted so that Ipk turn-off is achieved at 5.0 ms. 10 15 20 30
VCC = 20 V CT = 1.0 mF TA = 25C
0.15 0.1 7.0
10 k T 7.0
10
15
20
30
50
70
40
50
70
RT, TIMING RESISTOR (k)
RT, TIMING RESISTOR (k)
Figure 2. Oscillator Frequency versus Timing Resistor
Figure 3. Power Switch Peak Drain Current versus Timing Resistor
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A snubber is required across the primary winding because of the leakage inductance. An RCD snubber is implemented. The diode will be an MURS160 ultrafast diode. The resistor and capacitor are dependent upon the leakage inductance of the primary windings of the transformer. Preliminary values of the resistor and capacitor are derived from the following equations. The final values will be determined on the bench with an examination of the trade-offs between system efficiency and peak voltage.
Vreflect + (Vout ) Vdiode) * R+ Np Ns (eq. 10)
Np
Feedback Network
2 * Vmax * (Vmax * (Vout ) Vdiode) * Ns ) Lleak * Ip2 * freq (eq. 11) C+ Vmax Vcr * R * freq (eq. 12)
See AND8023/D page 7 for further information on equations 11 and 12.(2)
Vmax is the desired peak voltage on the power switch minus the bulk voltage, Vcr is the clamping ripple usually 20 V, and freq is the switching frequency of the MC33363B. Most of the values necessary to calculate the secondary components are complete. To select the diode, the secondary current peak, the blocking voltage and the forward voltage drop are the main criteria. Ripple current from the transformer, the output voltage and the output ripple current are the criteria to select the output capacitors.
Vblock + Vout ) Vdc * Ns Np C+ 1 Vripple * freq (eq. 13) (eq. 14)
With the low voltage output requirement, a programmable precision reference with a low operating voltage is essential. Isolated feedback networks provide an additional challenge since optoisolator diode forward voltages remain at 1.25 volts. With a conventional circuit using the TL431 and TLV431, the lowest ideal power supply output voltages are 3.75 and 2.5 volts respectively. If the reference operating voltage range is reduced to 0.9 volts, the minimum output voltage is lowered to 2.15 volts. See Figure 4 for the circuit schematic. This reduction is significant but will not satisfy a 1.8 volt output. By adding a resistor and PNP transistor, the reference voltage is reduced to the emitter-base voltage drop plus the programmable reference operating voltage. With the conventional TLV431 the minimum operating voltage is reduced to 1.95 volts. With the reduced reference voltage available in the NCP100, the minimum operating voltage is 1.6 volts. See Figure 5 for the circuit schematic. Figure 6 illustrates the output voltage ranges between the TL431, TLV431 and NCP100.
Vout R3 R1 U2 C2 U1 R2 GND
Utilizing the above equations and previous calculations, the diode needs to be rated at a minimum of 13 volts and 4.44 amperes. The MBRD835L is selected as the Schottky diode. The typical forward diode drop is 0.40 volts at 4.0 amperes. Rubycon capacitor YXG 6.3 volts 820 uF is chosen for the output filter. The voltage on the output capacitor needs to be rated twice the output voltage for safety. The rated maximum allowable ripple current of the capacitors is 0.865 amperes, so 2 capacitors in parallel will be used. A 1.0 uF ceramic capacitor will also be used in parallel with the aluminum electrolytics for fast transient response. Equations 13 and 14 are used to calculate the auxiliary winding values. The auxiliary winding diode is an ultrafast recovery surface mount, MURS120. The diode is rated for 1 ampere and 200 volts blocking. The typical forward voltage drop is 0.6 volts. The higher forward voltage drop is acceptable due to the lower power requirements of the auxiliary versus the secondary output. The auxiliary winding filter is the capacitor on the VCC pin.
Figure 4. Traditional Isolated Feedback Network
Vout R3 Q1 C2 R1
R4 C3 U2 U1 NCP100
R2 GND
Figure 5. 1.8 Volt Isolated Feedback Network
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36 16 Extended Range with PNP
6.0 3.75 3.2 2.5 1.95 TLV431 2.15 1.6 NCP100
Figure 5 illustrates an additional 1.0 uF capacitor, C3, which is required for normal operation of the NCP100. Bandwidth concerns are raised when a capacitor is placed from the cathode to anode of the precision reference. Since the PNP transistor is used to drive the optocoupler, the cathode current of the NCP100 is amplified by the beta of the transistor therefore minimizing the effects of C3. R3 is a factor in the overall gain of the feedback network and R4 limits the current into the diode of the optocoupler. The roll-off frequency of the feedback network is related to R1 and C2 only. Figure 7 shows the circuit schematic, the gain and the phase of the TLV431 and the NCP100 with and without the PNP.
Power Supply Ideal Output Voltage Range (V)
TL431
Figure 6. Power Supply Output Voltage Range Comparison for Programmable Precision References
Vout ROSC ROSC Vout
R3
R1 Q1 C2
R3
R1
C2 C3 R2 GND R4 U1 R2 GND
C3 U1
Figure 7. Schematics for Gain and Phase Comparison of TLV431 and NCP100 Note: C3 only for NCP100
180 40 70 GAIN (dB) 60 50 40 Gain 90 PHASE () GAIN (dB) 0 -90 -180 30 20 Phase 10 0 -10 -20 -30 Gain Phase
180 90 0 -90 -180 PHASE ()
10
100
1k Frequency (Hz)
10 k
100 k
10
100
1k Frequency (Hz)
10 k
100 k
Figure 8. TLV431 with PNP
Figure 9. TLV431 without PNP
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60 50 GAIN (dB) 40 30 20 10 10 Gain 180 0 90 -10 PHASE () Phase 0 -90 -180 -40 100 1k Frequency (Hz) 10 k 100 k 10 100 1k Frequency (Hz) 10 k 100 k 0 -20 -30 -180 Phase -90 PHASE () GAIN (dB) Gain 90 180
Figure 10. NCP100 with PNP
Figure 11. NCP100 without PNP
The values for the components in Figure 7 are as follows. Rosc = 12 Ohms, R1 = 12 k Ohms, R2 is adjustable dependant upon U1, R3 and R4 = 100 Ohms, C2 = 0.01 uF, and C3 = 1.0 uF. The open loop gain and phase test injects the oscillator signal across Rosc. The reference voltage is measured between Rosc and R1 to ground. The test voltage is measured at the collector of Q1 for the PNP circuit or at the cathode of U1 for the traditional circuit. Vout is set to 4.0 volts and R2 is adjusted so the DC test voltage is at the center of its range. R2 is the only component adjusted since it is neglected for AC analysis. There are negligible differences for the open loop gain and phase of the NCP100 with C3 compared to the TLV431 without C3 in the PNP circuit. Therefore there is no penalty with the added C3 in the NCP100 circuit. If the PNP transistor is removed, the TLV431 open loop responses with and without C3 are very similar with a single pole roll-off. The pole appears to be near 4 Hz. The NCP100 without the PNP has flat gain until a pole at 1.3 kHz due to R1 and C2. The plot remains at
-20 dB/decade and -90 degree phase margin until well beyond 100 kHz. The overall gain of the feedback network can be limited due to R1, R3 and C2 in Figure 7 . Varying these components will modify the maximum gain of the system. Since the NCP100 provides a single pole roll-off, the MC33363B compensation pin will not provide this function. The feedback pin of the MC33363B is connected directly to the reference pin. This will keep the output of the error amp low. A diode is in series with the output of the op-amp allowing the compensation pin to directly control the feedback to the oscillator ramp. The compensation pin is connected to the feedback pin via a 2.7 k Ohm resistor. The collector of the optocoupler is also connected to the compensation pin and the emitter is grounded. This completes the design of the system. See Figure 14 for the board layout of the schematic in Figure 1 and Table 1 for the component values.
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AND8031/D
Results
Table A lists the system parameters. Figure A illustrates the single pole feedback from the circuit in Figure 1 and the layout in Figure 13. The output current step response has a negligible effect on the output voltage. Figure B is the output
step response of the circuit from 20% to 100%. The circuit performance achieves the desired results for the off-line converter. The efficiency is low, but is expected for a low power off-line converter.
20 Gain 0 -20 -40 Phase
180 90 0 -90 -180 PHASE ()
10 Hz
100 Hz
1 kHz
10 kHz
Figure 12. Gain and phase results from the 1.8 volt 1 ampere off-line converter.
Figure 13. Output step response from 0.2 to 1.0 Amperes.
Table 1. System Parameters
Parameter Operating Frequency Primary Peak Current Limit Efficiency at 1.0 Ampere Load, 120 VACin Efficiency at 1.0 Ampere Load, 230 VACin Load Regulation, 120 VACin, 0.1 to 1.0 Ampere Load Line Regulation, 1.0 Ampere Load, 90 to 265 VACin Low Frequency Output Ripple, 1.0 Ampere Load High Frequency Output Ripple, 1.0 Ampere Load Value 110 kHz 230 mA 58.6 % 57.2 % 22 mV 1 mV 43 mVpp 350 mVpp
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Figure 14. Layout of Off Line Power Supply
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Table 2. Component Values of Off-Line Power Supply
Reference C1 C2 C3 C4,C10,C12 C5 C6 C7,C8 C9,C11 D1,D2,D3,D4 D5 D6 D7 JP1,JP2 Q1 R1,R2 R3 R4 R6 R9 R10 R11 R12 U1 U2 U3 U4 BC858ALT1 4.7 30 k 3.9 k 2.7 k 12 k 7.5 k 10 120 MC33363BDW Transformer SFH615-4 NCP100 Part 1n 4.7 u 390 p 1u 47 p 10 u 820 u 0.1 u 1N4005 MURS160T3 MURS120T3 MBRD835L Description X Capacitor 400 Volt Bulk YK Series SMT Ceramic 0805 SMT Ceramic 1206 1 k Volt Ceramic Disc 25 Volt MS5 Series 6.3 Volt YXG Series Y Capacitor Standard Recovery Rectifiers 600 Volt Ultra Fast Rectifier 200 Volt Ultra Fast Rectifier 35 Volt Schottky Rectifier 3.5 mm 2 Pole Connector Small Signal PNP SMT 0805 SMT 0805 1/4 Watt SMT 0805 SMT 1206 SMT 0805 SMT 0805 SMT 0805 SMPS Controller with FET Transformer EE16 CTX22-14966 4 pin Optocoupler Programmable Precision Reference ON Semiconductor ON Semiconductor Coiltronics Rubycon Rubycon TDK ON Semiconductor ON Semiconductor ON Semiconductor ON Semiconductor Wieland ON Semiconductor Rubycon TDK TDK Manufacturer
References 1. Brown, Marty, Power Supply Cookbook, Newton, MA, Butterworth-Heinemann, 1994. 2. Basso, Christophe, Application Note: "AND8023: Implementing the NCP1200 in Low-Cost AC/DC Converters,'' ON Semiconductor, August, 2000.
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Notes
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Notes
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Notes
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